When designing a receiver that is intended to be used for
optical through-the-air communications, there are some
things to be considered that are very different from what
might be required in most other situations where detection
(and demodulation) of an optical signal is to be done:
- Weaker signals. Signals are likely weaker in
through-the-air communications - typically by several orders
of magnitude - than in more familiar situations where an
optical signal is to be detected from optical fiber systems
such as optical fibers, TOSLINK, infrared TV remote, etc. In
thse situations the energy present at the receiver is really
quite high, suffering only modest attenuation through the
medium. Optical cables tend to have quite low loss,
while infrared remotes are usually pointed directly at the
device being controlled.
- Lower bandwidth. Effective bandwidth is
usually lower in long-distance through-the-air
communications. For weak-signal detection in
through-the-air optical communications and experimentation,
voice bandwidth (up to 3 kHz or so) is usually all that is
required, while specialized communications techniques
(carrier detection, WSJT and WOLF modes) work well with
frequencies below 300 Hz. If extreme weak-signal and
distance operation is not required, it is possible to use
subcarriers into the 10's of kHz and achieve reasonable
results.
What this means is that techniques for detecting
higher-bandwidth signals (IR remote controls, fiber optic
receivers, etc.) are not particularly well-suited for the
detection of weak, through-the-air signals.
Additional comments
about this page:
- Special emphasis is given to weak signal reception using
inexpensive, easily reproducible equipment that can be
constructed by a hobbyist.
- This page deals exclusively with circuits using
photodiodes for detection of optical signals and is
specifically targeted toward highly sensitive, low bandwidth
detection schemes - that is, below 3 kHz.
- Note that the detectors here are radiometric - that is,
they simply detect any optical energy that might strike them
without particular regard to the wavelength of light.
Note that in some cases, filters are added, but the
detectors themselves are generally
wavelength-insensitive.)
- This page is not intended as an in-depth
treatise on the theory and operation of optical detection
schemes.
- For more information on various optical
detectors it is strongly recommended that one
also read "Modulated
Light
DX Receiver Circuitry"
web page on the Modulated
Light DX site - an
excellent collection of pages compiled by Mike Groth and
Chris Long. At the bottom of this page is a
reproduction of an informative application note dealing
specifically with the use of photodiodes that is a "must
read."
Different types of
detectors:
Vacuum tube detectors:
Among the older types of detectors, one is the
phototube,
and another is its far more sensitive cousin, the
photomultiplier
tube (PMT). Still used today, the
photomultiplier tube is unmatched in its ability to detect
extremely low levels of light (especially at shorter
wavelengths) with fast response - provided that the tube's
spectral sensitivity curve is well-matched to the wavelength of
light of interest.
A quick examination of photomultiplier specifications
(such
information may be found about halfway through the Optical
Communications
for the Radio Amateur article by Chris Long) will
indicate
that
most
devices
are
fairly
insensitive
toward
the
"red" end of the spectrum: The so-called red-sensitive
"S-1" types have fairly miserable (0.1%) quantum efficiency in
the red wavelengths while more common S-4 types have quantum
efficiencies around one percent in the 600-650 nanometer
area. While there are some good, red-sensitive devices
available (such as the Burle C31034 series with a GaAs
photocathodes or some of the Hamamatsu multialkalai devices such
as those in the R669 or R7400 series) these tend to be fairly
expensive when bought as new devices and rarely show up on the
surplus market. Even the garden variety photomultiplier
tubes that are available on the surplus market (such as the
venerable 931A types) tend to be much more expensive than a
simple photodiode.
Photomultiplier tubes are also somewhat difficult to use:
They typically require 800-1500 volts for operation (depending
on the tube and application) and its large target area makes it
slightly more awkward to optimally illuminate with very simple
optics. Also to consider is that they are extremely
fragile, both physically - because they have fragile glass
envelopes - and, especially, optically: A good
photomultiplier can be wrecked by even a brief exposure to
daylight or strong light sources. Depending on the
particular tube, the power supply conditions and the intensity
of light, this "damaging" effect may only be temporary, with the
tube returning to "normal" in a few hours or days, or permanent
damage may have resulted in the diminution of the tube's
ultimate sensitivity and noise characteristics. Whether
the effects are temporary or not, a simple mishap could easily
end (or delay) experiments that are underway.
Note: In the future, I hope to run some
"A/B" comparison tests between the PIN diode optical receivers
and readily-available photomultipliers, such as the 931A and a
more modern multialkalai PMT.
Photoresistors:
Photoresistors
(such as the Cadmium Sulfide, or CdS types) are photoresistive -
that is, their resistance drops upon exposure to light.
While these can be fairly "sensitive" they are quite slow in
comparison to most other solid-state and vacuum tube devices -
on the order of minutes if one is looking at their
specifications for ultimate sensitivity (at very high
resistance) in near-total darkness. It is this extremely
slow response that makes them generally unsuitable for optical
through-the-air communications work, although they have been
used with limited success in short-range voice-bandwidth
communications systems where light levels are fairly high.
Another important factor is that the sensitivity of these types
of cells is mostly in the green visual wavelength - a distinct
disadvantage if one anticipates using red or infrared
wavelengths to minimize atmospheric effects.
Phototransistors:
Phototransistors
are convenient to use in that they are inherently
self-amplifying and can provide relatively high signal output
levels, but they do not have the ultimate sensitivity of
photodiodes. The sensitivity of phototransistors is
limited by their high intrinsic noise, much of which is a result
of collector-base leakage currents, and it is these noise
currents that tend swamp out the much weaker, photon-induced
currents at very low light levels. The small photoactive
area of typical phototransistors limits the amount of light that
they intercept (and thus sensitivity) if used without external
lenses and can make proper focusing of the distant light source
more difficult when used with lenses.
Photovoltaic cells:
Also called "
Solar Cells"
these are designed to produce electricity when exposed to
light. As detectors, however, they have a fairly slow
response and fairly high leakage current and capacitance - all
being distinct disadvantages when trying to use them to detect
very weak, modulated signals.
Photodiodes:
Photodiodes
are essentially very small photovoltaic ("solar") cells, but are
typically much smaller in area to minimize the capacitance and
they are optimized in their manufacture to minimize leakage
currents, intrinsic noise and to provide consistency amongst
devices. When photons hit the surface of a photodiode
electrons are mobilized producing currents that are proportional
to the amount of light. Photodiodes can also be operated
in a photoresistive mode in which the impingement of photons
results in current flowing through a reverse-biased photodiode.
Photodiodes are available in a large variety of sizes, from the
so-called "Small Area" types to the "Large Area" types. As
the name implies, the primary difference between these is the
actual size of the silicon substrate and the larger the
substrate, the higher the device capacitance. The
so-called Small Area photodiodes (typically one square
millimeter or smaller - sometimes
much smaller!) are
most often used in high-speed receivers: Their low
capacitance (typically well below 10pF) allows better frequency
response to be obtained. Large Area photodiodes (which can
be well over 10mm square) can "capture" more photons over their
surface area, but their response time slowed by their much
higher capacitance (in the 100's or 1000's of pF) so they are
often used where their larger area is desirable to accumulate
more photons - but speed isn't as important. It should be
noted that, when used with lenses, it is best that a
smaller-size diode be used, but with an area that exceeds the
diameter of the
"airy disk"
(assuming that the angular size of the distant light source is,
for all practical purposes, infinitely small) so that all of the
light being focused hits the silicon and can do some useful
work. In most cases it is not the airy disk (diffraction
limit) that dictates the minimum size of the detector, but the
"blur circle", the actual minimum size of spot of which the lens
system is capable due to imperfections of the (various) lens
element(s).
The optical response of silicon photodiodes is best in the
near-infrared wavelengths around 850-900 nanometers, but it is
still pretty good into the red portion of the spectrum, falling
off rapidly at shorter wavelengths. Fortunately, their
response is a reasonable match for optical through-the-air
communications involving red and/or near-infrared wavelengths -
the very wavelengths of interest in through-the-air optical
communications. The BPW34, for example, is typical of
silicon-based detectors in that it has a peak quantum efficiency
of 0.9 electrons/photon at 850 nm, but it drops to about 0.6
photons/electron at 630nm (the approximate wavelength of typical
high-power Red LEDs such as the Luxeons), to 0.4
photons/electron at 530nm (green) and down to around 0.25
photons/electron at 460nm (blue.) Some diodes have
wavelength-specific packaging to limit their response such as
the black (or darkly tinted) "infrared-only" versions that are
commonly used for infrared remote controls while others have
modified silicon structures and are manufactured differently to
enhance the green/blue wavelengths.
Another type of photodiode is the
Avalanche
Photodiode or APD. This device is
somewhat analogous to the photomultiplier tube in that it has
intrinsic amplification and has replaced the photomultiplier
tube in many applications, but the old photomultiplier
technology still wins when it comes to the ultimate in photon
sensitivity in many cases. Like the photomultiplier, the
APD requires a high voltage supply, but the requirements are
usually more modest, typically in the region of 100-500
volts. When operated near their maximum sensitivity their
use is somewhat complicated by the fact that several precautions
need to be taken in the design of the voltage source to assure
proper performance over varying operating conditions, arguably
making them more difficult to use than photomultiplier
tubes. At present, these devices are rather specialized
and are rather expensive when purchased new, are difficult to
find as "raw" devices on the surplus market, and are often
packaged as complete detector units that include the power
supply and amplifier. APDs are most useful where fairly
low light levels are encountered but high speed is needed.
Recently, testing was done using a receiver that uses APDs -
see the link at the bottom of the page.
Additional comments on appropriately sizing photodiodes:
As mentioned above, when used without optics, a larger
photodiode will intercept more photons than a smaller photodiode
because there is simply a greater "capture area" to be struck by
the photons. When used with external optics, however, the
size of the photodiode may be less-important in terms of
ultimate sensitivity because the light-gathering aperture is no
longer just the surface area of the photodiode itself, but the
capture area of the optics being used
with the
photodiode.
If you are using external optics to focus light onto a
photodiode's active surface - such as with a radiometric optical
receiver of the sort described on these pages - the size of the
photodiode is somewhat less important in terms of
sensitivity. What is most important is that not only is
the photodiode placed at the focus of the optics being used, but
that the area of the photodiode is somewhat larger than the
"airy disk" or "blur circle"
(see above) of the lens
system being used at optimal focus so that all light from the
intended source will, in fact, strike the active area of the
photodiode. If too small a photodiode is used then some of
the received light may be "wasted" - that is, spill out around
the photodiode and
not have its photons do the intended
job - that is, making electrons move about!
If a photodiode is used that has a much
larger
active area than the area of the "blur circle" of the light
focused onto it, several things happen:
- Excess "beamwidth". Off-axis light sources
can impinge on the photodiode, reducing the signal-noise
ratio of the source to be detected or, in other words, other
light sources near the distant one may also be detected and
"dilute" the desired signal.
- Contribution of noise by "unnecessary" detector
area: Much of the photoactive surface (that
which is not being illuminated by the distance
source) is "wasted" doing nothing at all except contributing
extra noise, further drowning out the desired signal.
(Remember: "X" number of photons will only excite
"Y" number of electrons, no matter how large your
photodiode!) Remember that the intrinsic noise
generated by the photodiode is generally proportional to
the area of the photodiode: The larger the diode,
the more noise it may contribute!
- Higher capacitance: A larger photodiode
necessarily has a higher capacitance: If you use a
diode that is excessively large, you are reducing the
bandwidth of the detection system because of this
capacitance!
What size of photodiode should be used? This question is
one that can be answered appropriately by knowing the
characteristics of your optics. Very high-quality glass
lenses should be capable of resolving a distant point of light
and focusing it onto a very small area, making the use of a
small-area photodiode quite practical. More imprecise
optics - such as Fresnel Lenses or less-precise plastic or glass
"conventional" lenses will have a larger "blur circle."
One important fact to recognize is practicality in actual
use: While extremely precise, finely-focused optics may
offer the best match for small-area photodiodes, constructing
and subsequent aiming them in the field will be correspondingly
more difficult. Unless it is the highest possible speed
that you are after it may, in fact, be more convenient to use
somewhat larger-area photodiodes than those that might be
optimally-matched to the size of the blur circle of your lens
(and allowing some "fudge factor" in system accuracy) to
increase the "spot size" on the diode. If this is done,
the ultimate sensitivity will suffer minimally (provided that
the larger photodiode's noise and capacitance characteristics
aren't the limiting factor) as all of the intercepted light is
still impinging on a photoactive surface, but aiming tolerances
may be relaxed somewhat, simplifying setup and potentially
improving long-term system stability.
For more detailed information on photodiodes,
read the application note at the bottom of the "Modulated
Light DX Receiver Circuitry" page.
A good starting point - the K3PGP receiver:
Figure 1:
This is a very sensitive optical receiver designed by
K3PGP. While extremely sensitive, it has rather
limited bandwidth. The version shown is suitable
only for nighttime use.
Click on the image for the same-sized version.
|
Let us first discuss one of the simplest possible "high
sensitivity" optical detectors - the so-called
K3PGP receiver
which shown in
Figure 1. While the receiver's
circuitry is simple, its actual operation is deceptively
complex.
One of the most striking aspects of this receiver is the
connection between the gate of the MPF102 and the
photodiode: If ideal component models were used, this
would simply be a floating junction and as the photodiode
reached full potential, charging would simply stop and the
circuit would not function - but real-world physical effects
come into play.
In this circuit the photodiode is operating as a photovoltaic
cell at very low light levels: As photons strike the
photodiode electrons are mobilized and a voltage appears at the
gate of Q1, the JFET, and change its conductance. Because
both the JFET and photodiode exhibit some leakage, this charge
will drain away and the voltage on the gate of the JFET will
eventually reach equilibrium and be more-or-less proportional to
the amount of light hitting the photodiode.
Assuming that a typical "medium area" photodiode is used (like a
BPW34 - a fairly good, but inexpensive device) the capacitance
of the photodiode will be in the general area of 70-80pF.
While this may not sound like much capacitance, this is, in
fact, enough to severely limit the frequency response to only a
few hundred Hz: After building and testing this circuit I
observed that the -6dB rolloff point, using a BPW34 diode, was
around 200 Hz under very low light conditions.
This rolloff is due largely to the capacitance of the photodiode
(about 75 pF in the case of the BPW34) being paralleled by a
high a resistance which is intrinsic to the photodiode and the
JFET, largely in the form of leakage currents. If one does
some simple math, it can be seen that this leakage resistance
could be modeled by paralleling an ideal JFET and photodiode
(e.g. those with no leakage currents of their own) with a
resistor in the range of 10 Megohms or so to simulate low-light
conditions. It should be pointed out that this is a very
incomplete analysis as other factors should be considered (e.g.
Miller
Effect of the JFET, photoconductive effects of the
photodiode - parameters that depend heavily on the amount of
light, etc.) but this very simple model will suffice for the
illustration of the frequency response limitation.
The rest of the circuit is fairly straightforward: The
JFET (Q1) forms a common-source amplifier providing significant
gain, while the following common-emitter bipolar stage (Q2)
provides even more gain. This circuit cannot tolerate very
much ambient light before the photodiode will achieve its
maximum open-circuit voltage and/or the JFET stage will
saturate, so it is most useful at
very low light
conditions and at low (<300 Hz) audio frequencies.
Comments:
- I constructed the K3PGP circuit using a 2N5457 for
the JFET and a 2N5089 transistor for the bipolar stage instead
of the MPF102 and 2N5088/2N4124 originally suggested.
Both of these devices are generally better-characterized in
terms of noise performance and other operating parameters than
the original devices although the builder should be prepared
to sort among individual devices if the goal is to construct a
circuit with the lowest-possible noise and best overall
sensitivity.
- Some of the transistors used for the circuits on this page are
getting increasingly difficult to find in "through-hole" package, but
are readily available in surface-mount. For example:
- The MMBF5457 is equivalent to the 2N5457.
- The MMBF102, BFT46 and MMBF4416 may be used in place of the MPF102.
- The SMD equivalent of the 2N5089 is the MMBT5089.
- The MMBT6428 is equivalent to the MPSA18.
For a more-detailed discussion of this circuit, see the "Modulated
Light DX Receiver Circuitry" page.
The VK7MJ optical receiver:
Figure 2:
The well-proven VK7MJ Optical receiver. Negative
feedback allows this to operate as a transimpedance
amplifier and improve bandwidth - but at the expense of
sensitivity.
Click on the image for a larger version.
|
Figure 2 shows the VK7MJ Optical receiver. This
well-proven design was devised by Mike Groth, VK7MJ, having been
adapted from circuits used to detect low-level emissions in
nuclear medicine.
Before we get to the photodiode portion, let's examine the
amplifying portion of this circuit: The input JFET, a
2N5457, is wired with the BC179 as a cascode amplifier.
This configuration greatly reduces the Miller Effect (where the
gate-drain capacitance of Q1 is multiplied due to the voltage
swing of its drain) by having BC179 respond to the varying drain
current of the JFET instead - and the cascode configuration
provides a significant amount of gain as well. The output
of the BC179 is buffered by the BC109, wired as a high-impedance
bootstrap circuit, which is further buffered prior to the
output, by a source-follower circuit using an MPF102.
The biggest difference between this and the original K3PGP
circuit is the addition of a negative feedback path from the
output to the input. The addition of this path creates a
Transimpedance
amplifier - that is, the amplifier to responds
mostly to the
current being output by the photodiode
rather than the voltage and in doing this the swamping effects
of the capacitance on a changing voltage are effectively
reduced: Any voltage change from the photodiode is
amplified and countered by a sample of the inverted output fed
back into the photodiode-JFET gate junction through Rf, the
feedback resistor. To maintain amplifier stability a small
amount of feedback capacitance, Cf, is added to counter the
photodiode's own capacitance.
By virtue of this (mostly) "current-only" response the frequency
response of this circuit can be much better than the K3PGP
circuit in
Figure 1, but this improvement in frequency
response is not without costs: The addition of the
feedback circuit and the effectively paralleled resistances
decrease the sensitivity of this receiver as compared with the
K3PGP circuit, not only by the addition of noise sources from
the added components, but by a reduction of the amount of signal
from the photodiode itself, further dropping already-weak
photon-induced currents farther down into the noise floor of the
active and passive components.
An additional feature of the VK7MJ circuit is the application of
reverse bias on the photodiode. In this circuit, about 5
volts of reverse bias is applied, effectively reducing the
photodiode's capacitance from around 75pF (for an unbiased
diode) to something in the area of 20-30pF. This
capacitance reduction has the expected effect of improving the
bandwidth, thus reducing the required amount of negative
feedback that would be required to accomplish the same amount of
bandwidth improvement, thereby improving the amplifier's
low-noise performance - particularly at higher
frequencies. One caveat of the addition of reverse bias is
that it has the potential to increase
Shot noise
(among other noise sources) due to leakage currents
through the diode - but this is a rather minor penalty at voice
frequencies, as it turns out, and only seems to be a significant
factor at very low (<200 Hz) audio frequencies.
Note that noise performance and gain may be improved by
increasing the value of Rf (consisting of R3 and R4 on the
schematic) the feedback resistor - at the cost of a reduction of
bandwidth. This particular circuit does not have
sufficient gain to allow effective use of a feedback resistor of
more than 50-60 megohms so further increases beyond this
resistance will not necessarily improve performance, but below
this effective "gain limit" imposed by the maximum value of Rf
(and the noise floor of the devices) that S/N will
increase. As noted in a message board comment by Yves,
F1AVY, increasing Rf from 10 Megohms to 40 Meg will cause a
four-fold increase in signal, but only a doubling of the noise
resulting in a net doubling of the signal-noise voltage ratio -
and all of this is at the expense of reducing the bandwidth!
As can be seen in
Figure 2 there is another variation of
the circuit intended for daylight operation and this
modification allows the circuit to operate under higher ambient
light conditions by capacitively isolating the DC components of
the photodiode from the rest of the circuit. Were this AC
coupling not done, a combination of the increasing
photoconductivity of the photodiode (in response to the higher
light levels) and the higher photovoltaic output could saturate
the amplifier stages fairly easily, causing them to slam to a
power supply rail. The use of this "daylight" modification
does result in inferior nighttime performance as compared
to the DC-coupled circuit, mostly owing to the addition of
another 10 Megohm resistor across the photodiode: It
should be noted that this resistor causes further attenuation of
the photodiode's output (dropping it further into the JFET's
intrinsic noise level ) and is, itself, a potential source of
thermal noise. When used in daylight, however, it is
likely that the limiting factor for the apparent system
sensitivity will be the fact that the distant transmitter will
be in a sea of noise - also known as daylight!
Comment:
I constructed a version of this receiver using a
2N5457 for the JFET, a 2N5087 in lieu of the BC179, a 2N5089
for the BC109, and an MPF102 as the source follower. All
of these devices have equal or better performance
specifications than the ones suggested on the schematic:
It is this circuit that I use as my "standard" reference.
For a more-detailed discussion of this circuit, see the "Modulated
Light DX Receiver Circuitry" page.
Improving on the VK7MJ receiver circuit:
Figure 3:
Top: Schematic of the improved transimpedance
optical receiver, version 2.02.
Bottom: As-built prototype of the circuit
wired in "PIF" configuration.
Click on either image for a larger version.
|
|
While the VK7MJ receiver is a well-proven and solid design, it
occurred to me that there were several things that could be done
to eke a bit more noise performance out of it - as well as
making it a bit more versatile:
- Increase the JFET's operating current. As the
drain current of a JFET is increased, the small-signal noise
current is a smaller proportion of the device's total
current, thus reducing the device's effective noise
figure. (This often referred to as "bulk current
noise.")
- Rework the circuit to make it much more tolerant of
ambient light. The VK7MJ circuit, if used under
conditions where the ambient light is much more than
"completely dark" tends to require readjustment of the bias
setting - a potentially finicky procedure.
- Make the adjustment of the feedback network much easier
to manage and more versatile. Since the
frequency response and noise characteristics are somewhat
dependent on the nature of the feedback path, it would be
nice to allow relatively easy optimization of those
parameters.
- Allow a much broader range of supply voltages.
The VK7MJ circuit requires at least 12 volts to operate
effectively. It would be convenient if operation on a
single, 9 volt battery were possible as it is generally
necessary that the receiver be operated from its own,
independent power supply to avoid the inevitable interaction
that would otherwise occur between its highly-sensitive
circuitry and the other devices sharing the power supply.
The results of these modifications may be seen in the circuit
shown in
Figure 3.
As can be seen, the cascode arrangement is maintained with Q1
and Q2, but a significant difference is the addition of Q3, a
bipolar current source. Q3 is used to maintain a fairly
constant, high level bias current in Q1, the JFET, to minimize
its noise contribution, but because the Q3 current source
operates at a fairly high impedance, it can supply current to
the JFET without incurring signal losses that would otherwise
occur were a low ohmic value of resistance used to supply a
similar amount of current. A further advantage of the
current source is its ability to supply a fairly consistent
current over a wide range of supply voltages without the need of
significant readjustment.
The bipolar (Q2) section of the cascode arrangement operates at
a comparatively low current and high impedance and by doing so
it can operate at fairly high gain without requiring
particularly high supply voltages. This cascode circuit is
somewhat unusual in that it is self-biasing: Because the
drain voltage of Q1 can vary depending on differing conditions
and with different devices, it is necessary to have Q2 maintain
a reasonably constant current contribution under all conditions
- but it is also essential that the AC base impedance be quite
low in order for it to have high signal gain, hence the R6/R7/C3
arrangement which allows Q2 to "track" Q1's DC properties.
The circuit R8/C4/U1b tracks the DC level from the cascode
circuit so that the audio amplifier, U1a, is bias properly under
a wide variety of operating conditions.
The remainder of the circuit consists of a simple noninverting
op amp gain stage that amplifies the high-impedance signals from
the collector of Q2 while minimally loading it: There is
nothing particularly special about this amplifier, except that
it should be of fairly a low noise type, but exotic amplifiers
need not be used. In this case, it is wired to provide a
voltage gain of about 33 - enough to provide enough source
signal for a feedback circuit, but the gain could easily be made
variable by substituting a potentiometer for either R12 or R13.
Comments:
- U1 should be a fairly low-noise amplifier.
The specified LM833 is an excellent choice, but the
more-common TL082, TL072, or LF353 will also work
well. A '1458 type amplifier may be usable in a pinch,
but it is demonstrably noisy by comparison!
- The value of R4 may need to be adjusted to suit the
characteristics of the JFET (Q1) that is used: A
typical range of values would be from 120 to 220 ohms.
If the value of R4 is too low (e.g. too much current) the
gain and noise performance will suffer. If it is too
high, the JFET may be running at a lower-than-optimal
current for the lowest possible noise performance. It
is better to err on the side of too-little current.
For proper operation, the drain-source voltage of Q1
will be well below 2 volts in total darkness under all bias
conditions.
- The values of the smaller electrolytic capacitors (below
10uF) are not critical - that is, a 4.7 or 10uF capacitor
can be used. The exception to this is C3, which should
be around 2.2uF to maintain circuit stability: Too low
a value and the AC performance suffers, while too-high a
value and the circuit can oscillate or ring (at low
frequencies) under some conditions - notably at higher
supply voltages and/or when there are sudden, wide
variations in light.
Adjusting for the proper amount of feedback:
As can be seen from bottom of the diagram, several
configurations are offered - and we want to apply the "
standard"
configuration of using a high-value feedback resistor. One
of the ways that this circuit differs from the VK7MJ circuit is
the way in which feedback is applied: There are provisions
to vary the amount of signal (using R10) being put into the
"hot" side of the feedback resistor and thus provide the exact
amount of feedback to obtain a flat frequency response.
This adjustment is approximately thus:
- The value of Rf is chosen. For a BPW34, a
value from 10 Megohms to 150 Megohms is appropriate,
depending on the ultimate bandwidth desired. Note:
Testing showed that there is relatively little performance
improvement gained by increasing Rf from around 50 Meg to
150 Meg as the main sensitivity limitation is the noise
contribution of the JFET and/or the photodiode. Note
that lower values of Rf will result in better toleration
of ambient light.
- Adjust for the critical amount of feedback:
Optically coupling a square-wave modulated LED (a frequency
of 1 kHz is a good value) into the photodiode, adjusting R10
for the "squarest" looking output, as viewed on an
oscilloscope, with minimal amount of overshoot and
overshoot.
- Important note: One must make certain
that the LED and optical detector are spaced at least 3
feet (1 meter) apart to prevent electrical
fields from the LED's square-wave generator and LED from
getting directly into the optical receiver: Verify
that the coupling is, in fact, optical only by
blocking the path with a piece of cardboard or other
opaque material.
- Also, use only the minimum amount of light
necessary to get a good reading on the
oscilloscope: Too much light will overdrive the
diode and cause misleading readings, making accurate and
repeatable adjustments more difficult.
- Alternate method of adjusting for proper feedback:
Use
a
sine
wave
generator
to
modulate
a
DC-biased
LED
(for
a linear brightness response) and then sweep the frequency,
from a few hundred hertz to where the amplitude rolloff
becomes dramatic. If the feedback is too high, there
will be a definite "peak" in the amplitude - or it may even
oscillate. If the feedback is too low, the frequency
response will roll off rather gradually. At the
critical amount of feedback, the frequency response will be
flat (or even exhibit a very slight peak - which is
usually OK) before it suddenly starts to roll off.
This procedure may be done either with a scope or with an AC
voltmeter that operates over the desired frequency range.
Potentiometer R11 is used to set the proper amount of gate
bias. For good operation, this is typically set at a
voltage that is roughly equal to (or slightly below) Q1's drain
voltage and this will often vary from one JFET to another so the
best voltage for lowest-noise operation (particularly with
feedback resistors above 50 Megohms) will have to be determined
by experiment. Perhaps the easiest method is to make the
adjustment in total darkness, but with a weak (very dim) optical
signal, adjusting R11 from one extreme where the receiver works
properly to the other, and then setting the potentiometer in the
middle of that range. Note that the bias voltage can be
tweaked somewhat to improve performance under conditions of high
ambient light.
It should be noted that with the addition of R10, the "feedback
adj" that the "Cf" (feedback capacitor) noted in
Figure 2
may not be required if R10 is adjusted properly, with the
intrinsic capacitance of the feedback resistor and other
components being adequate. It has been suggested that
slight improvements in performance may be possible with the
addition of a small amount of additional of feedback capacitance
(about 0.5pF to 2pF) and a reduced amount of feedback, but I did
not note any obvious performance advantage in doing so -
probably due to the presence of stray circuit capacitance.
If the circuit tends to oscillate or is excessively "peaky" in
terms of frequency response and adjustment of R10 doesn't seem
to help, try a larger amount of capacitance for Cf - but it is
unlikely that much more than 5pF would ever be required.
Improved ambient light tolerance:
One of the benefits of this circuit as compared to the original
VK7MJ circuit is that it is quite resistant to ambient light,
being able to tolerate wide variations without saturating:
This property makes this receiver a reasonable candidate for
"general" use in a wide variety of conditions - from total
darkness to an urban light-pollution setting - and maybe even
for "attenuated" daylight experimentation. Further testing
is required to see how the ambient light tolerance of this
circuit compares with that of the "daylight" version of the
VK7MJ circuit.
Operation over a wider supply voltage range:
Another advantage of this circuit design is that it operates
well over a wide voltage range - from 7 to over 14 volts,
drawing from 7 to 15 milliamps, depending on the voltage.
A caveat here: At differing supply voltages, not only does
the gain of the Q1/Q2/Q3 circuit change somewhat, but so does
the amount of reverse bias on the photodiode and these two
factors will affect the optimal setting of R10, the feedback
adjustment as well as Cf, if it is used. In practice, one
would set R10 at the voltage at which operation was expected,
but good performance could still be expected (albeit with a
somewhat different frequency response) at different
voltages. If a Zener diode (9 volts or so) is installed
from the base of Q4 to ground, this problem can be avoided if
the power supply is above 10-11 volts and regulation is
occurring.
It is important to be aware that your choice of op amp may also
be a limiting factor in how low of an operating voltage will
still yield good performance. I found that the circuit
still performed well (if R2's value was reduced as mentioned
below) at about 6 volts - even though this was below the
published supply voltage specification of the TL082 and LM833 op
amps that I tested. At these low voltages the gain of the
JFET/Bipolar circuit drops off noticeably and the reduction of
the photodiode's reverse bias causes frequency response to
suffer due to increased capacitance, both being factors that
require a readjustment of the feedback.
Comment:
It is recommended that resistor R2 in Figure 2
be reduced to 100 ohms or even 10 ohms if it has been
determined that a zero or slightly positive gate bias was
appropriate for the JFET used. Lowering the gate bias
would also allow for a commensurate increase in the reverse
bias of the D1, the photodiode as well as permit the circuit
to operate at a lower operating voltage.
Note about the "PIF" configuration:
In
Figure 3 (both on the schematic and in the caption)
there is mention of a "PIF" (
Photodiode
In
Feedback)
configuration.
This configuration is mentioned in the paper "Low-Noise
Photodiode Amplifier Circuit" by Hyyppa and Ericson
(IEEE
Journal of Solid-State Circuits, Vol. 29 No. 3, March 1994,
pp. 362-365) where a small amount of negative feedback was
applied to the "cold" end (e.g. the non-signal side) of the
photodiode. The claimed advantage of this circuit is that
it eliminates the need to establish a feedback path at the
junction of the photodiode and the gate of the JFET - a
potential noise source.
A copy of this article may
be found AT THIS WEB SITE. (Try this link if that doesn't work.)
As noted at the bottom of the schematic shown in
Figure 3
there is a mention of a circuit configuration to provide the
"PIF" circuit. When tested, it was observed that the PIF
circuit did, in fact, have a performance advantage over the
conventional "feedback" type circuit - but
only
below the "knee" frequency - that is, the frequency at which the
capacitance of the photodiode was causing a 6dB/octave rolloff
to occur: At frequencies much above this "knee" frequency
it appeared to offer no advantage as these AC components were
simply bypassed by the photodiode's own capacitance.
What this means is that in experiments using a "medium area"
photodiode like the BPW34, there was no performance advantage
above about 200 Hz. This has to do with the fact that at
the higher frequencies, the parallel capacitance of the
photodiode essentially bypasses the feedback, negating the
beneficial effects of the feedback. It is worth mentioning
that the photodiode mentioned in the article (the Siemens
SFH229) has a much smaller area than the BPW34 and the "knee"
frequency would be quite a bit higher than with the BPW34 owing
to the lower capacitance of that device. Although
experimentation with the "PIF" circuit didn't appear to have any
significant advantages for voice-bandwidth circuits when used
with the noted photodiodes, further experimentation may be
warranted.
Test setup to determine relative circuit
performance:
In order to provide a means of evaluating
relative
circuit performance I constructed a "Photon Range" in a basement
utility room without any windows. This test setup consists
simply of a red diffuse lens LED attached to the ceiling while
the circuit under test is placed on the floor (about 7 feet, or
about 2.1 meters) directly below it: In no case did the
LED or the receiver's photodiode have any optics. The LED
and receiver are connected via wire to an adjacent room and
using a function generator, the LED is driven with a square wave
and the current is set to just a few 10's of microamps -
just
enough to be able to perceive that the LED is illuminated at a
distance of several meters in total darkness with dark-adapted
eyes. The use of the generator allows the LED's modulation
frequency to be varied from less than 1 Hz to several megahertz,
although a frequencies above 10 kHz were not routinely used as
the computer's sound card's input frequency range was the
limiting factor.
For all testing it was verified that the noise floor being
observed was that of the receiver under test and
not
the noise floor of the computer's sound card! Typically,
the receiver's noise floor was at least 20dB greater than that
of the computer's sound card and any individual spectral
components from the test setup (usually related to pickup of
stray AC fields) were noted before the modulated optical source
(the LED on the ceiling) was activated.
The performance of the optical detector was measured by using a
laptop computer running the
Spectran program at a
bandwidth of 1.3 Hz. The signal-noise ratio was checked
using the same "standard" VK7MJ receiver - and the same unit was
used for all tests. For each test session, the first
and
last readings were done with this "standard" receiver not only
to verify that the equipment was configured the same as with
previous tests, but also to provide a basis for comparison for
the other tests and to make sure that the amount of light
emitted on the "Photon Range" was consistent during the entire
testing session.
In order to provide the best measurements it was usually
necessary to operate the laptop computer from battery to
minimize introduction of coupled AC line currents into the
receiver. Because these tests were done indoors, the
circuit under test was placed in a recessed, grounded metal box
(one with a top open to the LED mounted on the ceiling) to
provide shielding from stray AC fields that would have otherwise
ingressed the receiver, making measurements difficult - although
a bit of stray mains energy may be seen in some of the
plots: In those cases, the test frequency was chosen to
avoid proximity to that energy.
It should be noted that with the above test configuration one
should always use the same type (and size) of photodiodes to
draw relative performance comparisons. Without optics the
larger photodiodes will necessarily intercept more energy from
the light source than smaller ones, possibly causing an apparent
increase in the detected signal-noise ratio. If
different-sized photodiodes are used one should be prepared to
take into account the different areas of the devices used and
the subsequent amount of light that they would capture when
attempting to make direct comparisons between devices but, at
the same time, note that direct comparisons based solely on
device area is still subject to vagaries of the manufacturer's
specifications of "active" device area plus the fact that
different devices will have difference capacitances which will
affect the frequency response as well!
Figure 4:
Weak signal comparisons of the circuit of Figure 2 and
that of Figure 3. With Rf=22 Meg, the performance of
the VK7MJ and the "Version 2" receiver were identical.
Click on the image for a larger version.
|
Comparisons of receiver circuits:
Notes:
- In the accompanying signal-noise diagrams, the relative
signal-noise ratios are the only parameters that
should be considered as the absolute levels may have
varied. In all cases, the noise floor was that of
the receiver, not of the computer.
- Again, note that absolute signal levels may vary
between equipment test sessions. This explains why
readings in Figure 4 differ from those in Figure
6 given the same circuits. It is for this
reason that the VK7MJ circuit is always tested both at the
beginning and end of a testing session to assure a
reasonably consistent basis of relative comparison.
With the described test range, I
was able to quantify performance differences between the
various circuits, so I decided to test the sensitivity of
the VK7MJ circuit as compared to the circuit shown in Figure
3.
The first readings were done using the circuit in Figure
2 (Version 2.02) as a basis of comparison. For
those tests, I used a feedback resistor (Rf) with a value of
22 Megohms. I then checked the "Version 2" optical
receiver shown in Figure 3, also using a 22 Megohm
feedback resistor, and found the readings to be with a few
10ths of a dB of the VK7MJ circuit - too close to
call. In each case, the signal-noise ratio was
19.5-20.5dB, depending on frequency: A typical result
may be seen on the bottom row of Figure 6.
I then changed Rf to 54 Megohms in both receivers, making
modifications/adjustments to the feedback circuits as
necessary, and then re-ran the tests: The results are
shown in Figure 4. As can be seen, the two
circuits perform similarly - about 3dB better than with the
22 Meg feedback resistors - but the Figure 3 circuit
has a slight performance advantage over the original VK7MJ
circuit. This slight improvement is likely a result of
somewhat improved noise performance of the JFET input
stage's bias and amplification circuit as well as a slightly
lower noise contribution of the feedback circuit, but it is
also likely that some of it is due to normal variations in
the active devices being used.
After the test with 54 Megohms of feedback resistance, I
changed the Figure 3 circuit to use a 148 Megohm
feedback resistor. (The VK7MJ circuit did not have
sufficient gain to permit resistors higher than about 60
Meg to work properly.) I noted nearly identical
performance with the 148 Meg resistor as obtained with the
54 Meg resistor in terms of signal/noise ratio, indicating
that the sensitivity was likely being limited by the
performance of the photodiode and/or the JFET. I did
note, however, that with the 148 Meg feedback resistor, the
noise performance was more strongly affected by the setting
of R11, the bias resistor, than it was with a 54 Meg
feedback resistor, and that there seemed to be a wider
degree of component-related performance variation:
This has the implication that with the careful selection of
the lowest-noise components and the optimal setting of R11,
better performance may be obtained with the 148 Meg
resistance than with Rf=54 Meg.
Comment:
With a 54 Megohm feedback resistor, the
-1dB bandwidth of each receiver was about 30 kHz, but the
receiver in Figure 3 dropped to about 8 kHz or so
when Rf was increased to 148 Meg. Given the results
of these tests, there is likely to be little benefit of
using a feedback resistor of higher than 50-60 Meg unless
very careful component selection is made.
Evolution of another receiver circuit:
Having done these tests, I thought back again to the K3PGP
circuit (Figure 1) and wondered about its
performance. In my original tests I noted that while
this circuit provided excellent sensitivity, its frequency
response was, by itself, somewhat unusable for speech owing
to a 6dB/octave R/C rolloff that began at 200 Hz or
so. Despite the rolloff, I wondered what the
signal/noise ratio would be for a signal detected with this
circuit and, more importantly, if it would be better or
worse (and by how much) than that of the VK7MJ circuit in Figure
2.
Digging up my original K3PGP prototype, I did some tests
which yielded some interesting results, so repeated these
same tests using the Version 2.02 circuit in Figure 3
which I reconfigured to be without feedback or reverse bias
on the photodiode - essentially converting it into the K3PGP
circuit in that the photodiode was without bias or feedback
of any kind - and found that it slightly outperformed the
original K3PGP design, likely owing to the higher FET
current and the use of a cascode circuit - if not from
normal component variations. In these tests I observed
that while the signal output dropped off by 6dB per octave
(above the "knee" frequency of 200 Hz or so) the noise
dropped off at nearly the same rate! In
other words, the signal/noise ratio decreased at a slower
rate versus frequency than the amplitude did. At this
point I decided to apply reverse bias to the photodiode and
noted that higher frequency (>200 Hz) S/N and gain
performance improved markedly.
Figure 5:
Top: Optical receiver without feedback,
version 3.02 - see Figure 7 (below) for a simpler version.
Top center: Interior of enclosure with
version 3.02 circuit.
Bottom Center: Exterior of enclosure. A
strip of felt was used along the lid to prevent light
ingress between it and the body of the enclosure.
Bottom: The original prototype of the Version
3 circuit (e.g. no lowpass filter) as mounted in the
"cheap enclosure" (posterboard) optical transceiver.
Despite the lack of significant shielding, the circuit has
not proven to as susceptible to AC or RF fields as those
circuits using feedback.
Click on an image for a larger version.
|
|
|
|
With these promising results I constructed another
prototype, adding to it an op-amp differentiator to
compensate for the 6dB/octave rolloff caused by the
photodiode's capacitance and the lack of any feedback with
the ultimate result being the circuit shown in Figure 5.
Because
of the Q2/Q3 current source/cascode sections, the circuit is
able to operate properly even if the JFET (Q1) is conducting
heavily. It is because of this that the circuit still
works even when the photodiode is reverse-biased, causing
the gate-source voltage to become positive and even causing
the gate-source junction to conduct.
In many ways, the circuit operation is nearly identical to
that of the Version 2.02 circuit in Figure 3 - at
least until U1A. U1A is simply a unity-gain follower,
used to establish a low-impedance source for U1C, which is a
differentiator which has a classic 6dB/octave emphasis
curve that nicely cancels out most of the R/C rolloff of the
photodiode. As in the Version 2.02 circuit, the op
amp, U1, should be fairly low noise: I tried a TL074,
TL084, and LF347 with equally good results - all much
quieter than the Q1/Q2/Q3 amplifier, but an LM324 was
noticeably noisy, decreasing the the receiver's ultimate
sensitivity.
Added to this circuit is a 3.5 kHz lowpass filter that may
be switched in and out with S2 to remove some of the "hiss"
coming from the photodiode amplifier - the high frequency
components of which could cause "ear fatigue" when trying to
dig out signals with poor signal/noise ratios. The
lowpass filter also has the advantage that if an optical
signal is being received that is generated using PWM
techniques, the majority of the PWM switching components are
removed - an important consideration if you plan to record
the audio to a digital or magnetic take recorder or
computer, not to mention preventing a normal audio
amplifier from distorting from the PWM frequency
components. Note that the lowpass filter adds about 7
dB of audio gain. Also added is a gain switch (S1) -
just in case one is trying to detect a weak signal and one
needs as much audio as possible.
Comments:
- A simplified version of this same circuit may be seen in figure
8, below.
- A 9-10 volt Zener may be installed from the
base of Q4 to ground to regulate the voltage-sensitive
portions of this circuit if desired. This would be
appropriate when operating from a 12 volt supply that
might be shared with other equipment. In order to
prevent crosstalk it is strongly recommended that this
receiver be powered from its own, isolated power supply!
- If only digital operation is
anticipated, one may not need to bother with the
addition of the differentiator or lowpass filter at
all. It should be pointed out, however, that when
the suggested devices are used for U1, the noise
contribution that limits sensitivity is not U1,
but the photodiode front end (the Q1/Q2/Q3 circuit)
preceding it. It should be noted that some "sound
card mode" programs expect the audio response to be flat
and some artificial performance degradation may result
if the signals presented to it have a 6dB/octave
pre-differentiator "tilt" to them as might be seen from
pin 1 of U1.
- This circuit, like that shown in Figure 3,
has a wide range of operating voltage - from 6 to about
15 volts. Above 15 volts or so, the circuit
may become unstable due to excess gain in the Q1/Q2/Q3
circuit.
- When operating from a single 9 volt transistor
radio battery the current consumption is around 10-15
mA.
- TH1 (in Figure 5, top)
is a "self-resetting thermal fuse." The sole
purpose of this device is, along with D6, to prevent
damage if the power is applied with reverse
polarity. Because I have been powering this
receiver with a standard 9 volt alkaline transistor
radio battery, it is very easy to momentarily
reverse-connect the polarity while trying to connect the
battery in the dark - something that would likely
destroy U1, the op amp. If you do not use TH1, you
may substitute a 10 ohm, 1 watt resistor: This
resistor, along with D6, should be able to protect
against momentary shorts and because this circuit
consumes only 10-15 milliamps, the voltage drop across a
10 ohm resistor would be minimal. The use of a
normal fuse is not recommended as spares
would have to be kept onhand - something that is
difficult to do when trying to remember all of the other
gear that you need to bring along.
- This circuit is much less susceptible to
pickup of stray RF and AC noise fields than either the
VK7MJ circuit, or the on in figure 3 owing to
the lack of any other components connected at
the photodiode-JFET gate junction to intercept such
fields. As a result, less shielding is necessary
to obtain excellent results, provided that one use
minimum lead length between the photodiode and the
JFET. (I mount the JFET right at the
photodiode to minimize such effects.) The
optical receiver in my "cheap
enclosure" (see the bottom image in Figure
5) has minimal shielding (because it
was the prototype) and I had no problems with hum from
stray AC fields, even when the circuit was operated
indoors, nor were any RFI issued noted when it was
operated near a 2 meter amateur radio transceiver.
I would be reluctant to operate this
relatively-unshielded circuit on a site shared with high
power transmitters, however.
- As with the previous circuit, the value of the
bypass capacitor in the cascode amplifier (in this case,
C4 - 2.2uF) should be chosen with care: If very
large values are used, low-frequency oscillation or
"ringing" may occur, particularly at higher supply
voltages (e.g. >12 volts) or with large,
rapid/transient changes in light.
How it works:
It is worth mentioning the similarities and
differences between this, the K3PGP circuit, and a one using
a transimpedance amplifier - like the VK7MJ circuit:
- Like the K3PGP circuit, there is NO component connected to
the gate of the JFET (Q1) other than the photodiode.
- Important note: This connection should be
made in air and NOT on a circuit
board, as any leakage path - however slight, from
dirt, moisture or solder flux - can degrade performance.
- It is recommended that the JFET and photodiode be washed
clean with denatured alcohol to remove any residual solder
flux, dust or other contaiminants to prevent a leakage
path.
- The same high current/cascode circuit is used as in the
Version 2.02 circuit. One minor difference is that the
source resistor, R4, is a much lower value. Once
proper circuit operation has been determined, it should be
bypassed with a jumper, but its presence makes it easy to
measure Q1's source current.
- Unlike the K3PGP circuit, the photodiode has reverse
bias applied to it. As the graphs in Figure
6 show, the reverse bias has no significant negative
effect on performance at voice frequencies - at least at
very low levels of light. Only at very low frequencies
(well below 200 Hz) does the noise from the reverse bias of
the photodiode become apparent.
Operating a JFET with "gate current":
Perhaps the most unique aspect of this circuit is the fact that
the JFET's gate-source junction is, in fact, conducting!
From what I can tell, there are few (if any) other published
circuits that pre-date this one in which the gate of the JFET
being biased into conduction is an essential aspect of their
operation. Furthermore, there is surprisingly little
information to be found in the literature describing how JFETs
operate under conditions where gate current is flowing.
In my experimentation and by deriving curves I have observed that
the drain current of most depletion mode JFETs will continue to
increase even after the gate-source junction begins to conduct -
even to current levels well in excess of the saturation current
specified in the device's datasheet. As you might expect,
the gate-source voltage begins to follow the classic
voltage/current diode curve once gate-source conduction occurs.
Concerning this circuit configuration, some interesting things
happen:
- The photodiode itself becomes reverse-biased when the
JFET's gate rises about 0.6 volts above the source voltage
and the gate-source junction goes into conduction:
- The reverse-biasing of the photodiode, in addition to
everything else, reduces its capacitance and improves the
response at higher frequencies.
- With the JFET essentially at saturation, it is also
operating at its maximum current - something that is
conducive to its lowest-noise operation. I
find it somewhat surprising that this gate-source
conduction does not seem to be a major source
of noise in this circuit.
- As the photodiode current and gate current increases,
the impedance also decreases as the intrinsic gate-source
"diode" begins to conduct more. This effect is
minimal under dark conditions, however.
- The photodiode operates in its normal photovoltaic mode -
that is, it produces its own current when photons hit it.
- The photodiode also operates in the photoconductive mode -
that is, additional light will cause more electrons to flow
from through the diode from the bias source.
Particularly under high ambient light conditions (and higher
photodiode current) further increases in voltage across the
photodiode are somewhat prevented due to the low AC
impedance of the "cold" end of the photodiode (because of
the bypass capacitor) and the conductivity of the
gate-source junction on the "hot" end of the photodiode.
- At very low light levels the linearity of this circuit is
excellent - equal or better than that of the VK7MJ circuit -
and end-to-end distortion of an audio frequency optical link
(a modulated LED plus the receiver) was under 1%. Application
notes pertaining to the use of photodiodes (particularly
those published by Hamamatsu) indicate that the
application of reverse bias is, in fact, beneficial to
device linearity. At higher (room-ambient or
daylight) light levels distortion is much more apparent, but
has little impact on speech intelligibility.
- When using photodiodes with different device capacitances
- or even with different amounts of reverse bias on the same
photodiode - the "knee" frequency (that is, the frequency at
which the 6dB/octave rolloff begins to affect the bandwidth
of the photodiode circuit) will vary accordingly. If a
flat frequency response is required, adjustments of the
differentiator's components will be necessary, but for
typical speech operation this is usually unnecessary.
- Depending on the channel resistance of the JFET being
used, the drain-source voltage could be quite low! For
the MPF102 and 2N5457, a drain current in the 5-15 mA region
may yield a drain-source voltage anywhere from 0.5 to 2
volts. For a lower-resistance device like the BF862,
the drain voltage could be as low as 0.15-0.25 volts or so and
the circuit would still be functioning properly.
Remember that this is part of a cascode circuit and it
is not the voltage at the drain that carries
the signal, but variations in the drain
current!
Performance of the Version 3.02 circuit
Even before I did more scientific, comparative
testing in my "photon range," I could tell by ear that this
circuit easily outperformed any others that I had tried:
The results of comparative performance testing may be seen in
Figure
6.
Along the bottom row is the performance of the standard test
receiver, the VK7MJ circuit shown in
Figure 2. The
top row of
Figure 6 shows the performance of the circuit
in
Figure 5 when operated from an 11 volt supply - a
configuration that results in about 8.5 volts of reverse bias
across a BPW34 photodiode. As can be clearly seen,
the signal/noise ratio at 1250 Hz is about 14dB better than the
original VK7MJ circuit - and 8-9 dB better than the Version 2.02
circuit in
Figure 3. As expected, performance
degrades with higher frequency, but even at 5 kHz (the highest
frequency that I could test with my laptop) it was still
outperforming any other circuit that I had tried.
This circuit isn't without its drawbacks, though, as its
flat
high frequency response does have a distinct limit dictated by
the practical constraints of the differentiation circuit related
to the fact that the 6dB/octave increase cannot go on
indefinitely. While the "flat" audio afforded by the
circuitry is desirable for voice operation, it may not be
important when digital modes that are insensitive to "tilt"
(e.g. amplitude versus frequency) - which is true of most
narrowband digital signalling schemes. This lack of
"flatness" may be of also minimal importance if ultrasonic
subcarriers are uses: A dB or two of "un-flatness" across
the passband of an SSB signal at, say, 16 kHz, is unlikely to be
noticed!
For a more in-depth discussion on this
topic see the comments on the "Flat" audio output depicted
in Figure 8, below.
Figure 6:
Performance comparisons of the VK7MJ receiver shown in Figure
2 and the version 3.02 receiver shown in Figure
5.
Click on the image for a larger version.
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Notes about frequency response with the "Version
3.02" circuit:
Operation under conditions of high ambient light:
Unlike many other optical receivers such as the K3PGP, VK7MJ and
even the "non-daylight" Version 2 receivers which tend to "slam"
to a supply rail and go mute, the "Version 3" circuit will
operate to a degree even with extremely high light levels,
albeit with altered frequency response and distortion
characteristics as noted below. As far as is known, this
circuit is one of the few optical receivers that is useful all
the way from the smallest amount of light to full noonday sun
where it has thusfar been used to span in excess of 20
kilometers (about 12 miles) with good results!
One peculiar quirk of this circuit at higher levels of ambient
light is the fact that the frequency response becomes skewed and
the audio will begin to sound distinctly tinny. The reason
for this is that at higher levels of ambient light, the
photodiode becomes more and more conductive, and as this happens
the capacitance of the photodiode - the primary limitation of
high frequency response - is increasingly shunted by the lower
effective resistance, thereby improving high frequency
response. Additionally, as the photodiode current (along
with the gate current) increases, the gate impedance also drops,
further reducing the effects of the photodiode's
capacitance. Under very low-light conditions, the "knee"
frequency is around 150-250 Hz for the BPW34 (depending on the
photodiode's capacitance and the amount of reverse bias) and is
generally unnoticeable (except as a slight lack of "bass"
response) but with higher levels of light this "knee" moves well
into the middle of the audio range where the post-emphasis
effects of the differentiator become quite obvious in the audio.
Comment: It would be a fairly easy matter to
provide an extra control to adjust the "knee" frequency of the
differentiator to manually compensate for frequency response
differences as well as an extra resistor and capacitor to
recover the "lost" bass response - but the low-frequency
rolloff may be an advantage because of its tendency to
attenuate 100/120Hz hum from AC-powered light sources.
Under these conditions, one might consider taking the audio
from the "flat" output of the circuit depicted in Figure 8,
below.
With the increase of ambient light comes a dramatic increase in
noise as well - both from the photodiode itself (and possibly
the JFET) and the source of ambient light. While this
effect is, for all practical purposes, negligible in the voice
frequency range at very low light levels, it will eventually
become a roar of noise at much higher light levels - those at
and beyond the point where the audio becomes "tinny." For
this reason, if operating under conditions of high ambient light
or where the transmitting station is backgrounded by some light and/or a reflective background,
improved performance may result from adding a bit of optical
attenuation
to the receiver!
Remember: The receiver tested had about 14dB
better intrinsic sensitivity than the original VK7MJ receiver,
so you may very well be able to tolerate a bit of "optical
attenuation" on this circuit and still have performance that is
on par with the "non daylight" version of the VK7MJ
circuit. It is also worth noting that almost all (ambient)
light sources contain large amounts of thermal noise, so it may be that
the distant signal source may simply be being drowned in a sea
of optical noise from these other sources, anyway.
"No, the gate isn't
floating!"
At first glance of
Figures 5 and 8 it may appear that the JFET's
gate is floating: IT IS NOT!
Note that the "cold end" (non-gate side) of the
photodiode may be biased to a rather high voltage and
were the FET of an insulated gate
type the potential would try to rise to roughly match it
- at least until it broke down! Since it is
a junction FET, the "gate-drain diode" junction will
conduct and keep the "hot end" of the photodiode to
within about a "diode's drop" of the drain voltage which
- for most practical purposes - is at drain (ground)
potential.
This does several things:
- This allows a bias to be established across the
APD, both reducing its capacitance and allowing its
internal amplification properties to be realized.
- The FET is turned "on." As expected, the
channel resistance of the FET drops with increasing
gate-drain voltage but what is not commonly realized
is that with most JFETs, the channel resistance will
continue to decrease even as the gate-drain voltage
goes positive. Once the gate-drain junction
"diode" begins to conduct, the device's resistance
will continue to decrease as the voltage will still increase
although you now have a diode there with its
expected curve! If you are skeptical of this
observation, the construction of a simple test jig
using almost any common JFET will bear this out as
demonstrated in the graph below:
Figure 7:
Gate current versus drain current for a
typical JFET using measured values in a test
fixture.
Click on image for a larger version.
|
As can be seen in figure 7 the gate current increases
exponentially with gate-source voltage in a "diode-like"
manner. Like bipolar transistors, the drain
current (akin to collector current) increases
with gate current (akin to base current), but
it's in linear proportion to the gate voltage
rather than the gate current! This feature
is due to the fact that the "gate-source" junction is
conducting and is doing so in a classic "diode-like"
manner.
For our purposes the JFET operates in this mode in a
manner much more "quietly" than a bipolar transistor
would if we were to simply drop one in its place in this
circuit, mainly due to the fact that noise currents are
a small portion of the FET's overall drain current
whereas they would be comparatively large in the case of
a bipolar tranasistor.
Although this graph doesn't extend far enough, this
"semi bipolar-like" property of JFETs is exhibited only
for very low gate currents as the FET itself is "mostly"
saturated at the point that a significant amount of gate
current (e.g. gate current >> gate-source
leakage current) begins to flow and there is
a limit as to how much drain current will flow and still
exhibit any resemblance to the curve above!
Under low-light conditions, the operational and leakage
currents of the photodioe aren't enough to "saturate" the JFET
and it continues to operate "normally" - even with a
high (>100 volt) bias in a test receiver using an APD (Avalanche Photo Diode).
If operated under conditions with higher ambient (or
incident) light, the bias voltage should be reduced as
much as necessary and R202 will provide ample protection
to the photodiode (or APD) and FET to prevent either from being
damaged. It should be remembered that if there is
plenty of "extra" light, the extremely high sensitivity
of an photodiode or APD-based receiver isn't going to be required, anyway and
one might be better off using a different (and
less-sensitive) detector! |
If you intend to operate this circuit under high ambient light
conditions frequently it is recommended that one make the
photodiode bias variable. In referring to the Version 3.10
schematic in
Figure 7, the easiest way to do this would
be to put a potentiometer across the power supply and connect
the wiper to the (former) V+ end of R1. The value of this
potentiometer isn't critical, but something around 50k to 100k
would keep the static current drain of the potentiometer itself
to a level low enough that it won't significantly impact battery
life.
Expect the adjustment of this potentiometer to be somewhat
"touchy" at the low end, so if a linear taper potentiometer is
used, it is recommended that a fixed resistor of about 1/3 of
the potentiometer's total resistance be connected from the wiper
to the high (V+) and of the pot to "stretch" the low-end
adjustment a bit. If you have a potentiometer with a
logarithmic taper then simply wire it so that the "stretched"
portion (e.g. the least amount of resistance change per degree
of rotation) is at the low-voltage end of the bias. Note
that if the bias is set too low - especially under dark
conditions - it is possible that there will be too little drain
current in Q1 for the circuit to work and that there may be
NO
audio output at all.
Because the amount of bias affects the photodiodes capacitance
and - in bright light - affects the amount of conducted current,
the bias voltage will affect the frequency response. At
low light, a low bias voltage will make a very definite
difference in the "low end" audio response: Expect
somewhat similar things to happen under bright-light conditions.
Other circuit comments:
As can be seen from the schematic in
Figure 5 there are
provisions to add an external bias voltage. Under zero and
low-light conditions, the leakage currents of the photodiode are
probably in the nanoamp range - likely lower than the leakage of
the bypass capacitors - so one or two 9 volt transistor radio
batteries wired in series with the main supply may be used to
provide this voltage if it is desired - and an "off" switch for
the bias supply is likely unnecessary. Under "brighter"
conditions, the photodiode will conduct more heavily -
eventually being current limited by R1 and R2, but under normal
conditions, the amount of leakage experienced is likely to be a
small fraction of the self-discharge of the batteries that you
might use. As noted in
Figure 6, under low-light
conditions, however, a higher bias voltage (up to 30 volts) can
allow for further improvement in the signal-noise ratio at
higher audio frequencies (e.g. above 2 kHz or so.) If you
anticipate operating it under both low-light and "high-light"
conditions, you may wish to make the bias supply variable using
a potentiometer.
Figure 6 also shows some tests using a Hamamatsu S1223-01
photodiode - a larger, lower-leakage photodiode than the
BPW34. At lower frequencies, this device performs better -
largely because of its larger surface area (13mm
2 for
the S1223-01 versus 7.5mm
2 for the BPW34 ) allowing
it to accumulate more light (roughly 4.7dB more signal) in the
absence of optics - but at higher frequencies, its higher
capacitance begins to degrade performance.
Figure 6
nicely illustrating the limits of the efficacy of this circuit
at higher frequencies while providing a dramatic demonstration
of the improvement obtained by the lower junction capacitance
associated with higher reverse bias.
It is worth noting that the
very low (below
200 Hz) frequency performance may be hindered by the
application of reverse bias due to
"1/f" noise
- also known as
"Flicker
noise. In the case of
Figure 6,
this is shown by a slight degradation at 150 Hz when using the
S1223-01 photodiode - but this effect is more pronounced at
still-lower frequencies where the noise due to the reverse bias
leakage current has more impact. What this means is that
for
very low frequencies (in the 10's of Hz) it is
likely best to follow K3PGP's advice and to
not
apply reverse bias.
Theoretically, the S1223-01 should, when no optics are used,
have about 4.7dB better sensitivity than the BPW34 simply
because of its larger (1.7x) surface area - but this does not
take into account the fact that more surface area also means
more capacitance to swamp out signals and more photodiode
junction material to contribute noise (e.g. a higher "
NEP")
-
nor
does
it
necessarily
take
into
account
the
noise
from
the rest of the amplifier system following the diode.
When used with external optics, the size of the photodiode is
likely to be dictated more by how the distant light source is
focused onto the photosensitive material: In this case it
is best to use as small a photodiode as possible - provided that
the photodiode is at least as large as the "blur circle" of the
lens system being used and doing so minimizes photodiode
capacitance and leakage current - both of which improve the
signal/noise ratio. Additionally, a smaller-sized
photodetector can be used to reduce the beamwidth of the optical
receiver - something that can further improve the signal-noise
ratio by virtue of reducing the response to off-axis light
sources.
Simplified
version
of the "Version 3" optical receiver
Figure 8 shows a
simplified version of the "Version 3" optical receiver.
The performance of this circuit is the same as that shown in
Figure 5 (above) but a bit
of "minimizing" has been done - most notably the removal of the
power-supply filter (Q4) and the low-pass filter (U1D) as it is
expected that the receiver will
always be powered from
its own, independent power supply. Retained is the
high/low gain switch (S1) and the reverse-polarity protection
(D6 and TH1) as these items were considered to be important.
Also shown is a "Flat" audio output which precedes the
differentiator and may be useful for those wishing to experiment
with higher (>3 kHz) or lower (<300 Hz) modulated optical
signals. Note the presence of C11 (10uF) to block DC as
well as R15 (47 ohms) to provide a bit of protection against
capacitive loading - such as an audio cable - that could
de-stabilize U1a.
(Most op-amps are never happy with
large amounts of capacitance directly on their outputs!)
In this circuit, R4 is lowered to 10 ohms to reduce the voltage
drop and allow operation from lower supply voltage, and it is
used to measure the current through Q1, but this should be
bypassed with J1 after measurements are completed.
This circuit is intended to be operated from its own, single
9-volt battery - which is one of the reasons why the
reverse-polarity protection is present: It is extremely
easy to momentarily connect a 9-volt battery backwards while
fumbling in the dark - something that could instantly destroy
U1!
While the use of an LM833 has been shown, practically any
low-noise dual op-amp may be used. Note that operating an
LM833 from a single 9-volt battery pushes the low voltage limit
of this device which is 10 volts: Testing has indicated
that the LM833 seems to operate reasonably well down to at least
7 volts, but this is not a guaranteed specification! If
you are constructing this circuit keep in mind that there are
many other (perhaps less-common) op-amps that offer good
performance but can operate from much lower supply voltages,
such as the National LM4562 or the LMC6482.
Figure 8:
Simplified circuit of the "Version 3" optical receiver.
Click on the image for a larger version
|
Suitable enclosures and shielding:
The two center pictures in
Figure 5 show the enclosure,
constructed of double-sided copper-clad circuit board material,
containing the as-built version 3.02 circuit. It should be
noted that all signal and power leads are passed into and out of
the enclosure through solder-type feedthrough capacitors in
order to avoid the ingress of RF energy. A careful
observer will also note that in the center of the enclosure, set
back from the hole, one can see the photodiode and Q1, the
JFET: Note that the photodiode-to-gate connection is done
in midair to avoid any possible leakage paths that might occur
on circuit board material. Also, the photodiode is set
back from the hole by several millimeters to permit the
enclosure itself to provide some shielding of the photodiode
from any e-field energy that might be present. Finally,
note that the top of the inside of the enclosure is painted
black to minimize reflections from off-axis light sources that
might affect the sensitivity of the receiver.
As mentioned before, this circuit is less-susceptible to the
effects of stray AC and RF fields than either the VK7MJ or the
Version 2.02 circuit for one simple reason: The most
sensitive junction (that of the JFET's gate and photodiode) has
nothing else connected to it. In the case of the
other circuits, a feedback resistor is connected at this
most-sensitive junction and will more-readily pick up any stray
fields that may be present. Despite its relative immunity,
it is
still quite sensitive to AC fields so one should
still employ good ("VHF-style") construction practices when
building this circuit.
Thoughts on further performance enhancements:
The sensitivity performance of
the Version 3.x circuit is not to likely to be increased too
much, although some minor gains (a dB here and a dB there)
may be had from things like:
- The choice of a photodiode. Higher quality
photodiodes may have lower noise energy, as would
smaller-area photodiodes. It is important to note that
while smaller-area photodiodes may offer somewhat better
performance - especially in terms of speed - their small
area makes precise focusing of the received optical signal
onto a very small area more difficult: The area of a
BPW34 or similar diodes is a good compromise for speed and
the "spot size" that can readily be focused by typical high
quality molded plastic Fresnel lenses: If a smaller
photodiode is used additional optics may be used to further
reduce the spot size. Note also that with the lower
capacitance (and better high frequency response) it may be
necessary to modify the differentiator circuit to adjust the
"knee" frequency to maintain a flat frequency response
across the audio range.
- The choice of the JFET. The 2N5457 has
respectable noise performance, but there are other JFETs
that are rated for slightly lower noise. Another
possibility is to capitalize on the use of higher JFET
current to reduce noise by choosing a higher-current JFET
and/or paralleling multiple JFETs: The Philips BF862
appears to be a good candidate for testing. Another
way to improve performance is by paralleling FETs, but this
has the obvious caveat that doing so increases circuit
capacitances.
- JFET operating current, as set by resistor R5 (shown as
being 120 ohms). By appropriate selection of
this resistor, it is possible to "tune" the circuit for the
best noise performance for the particular transistor used at
Q1. For the 2N5457 shown, the value of 120 ohms is a
good starting point, while for the BF862 JFET, a lower
resistance (around 68 ohms) might be a good initial
value. Note that for many of these devices (Q1, the
current source Q3) are somewhat temperature-sensitive, so
the optimal current will vary somewhat. In
field-testing, however, no obvious temperature-related
effects were noted. When testing with random MPF102's,
a value of 220 ohms seemed like a good starting point.
- For many FETs (such as the 2N5457-2N5459 and the BF862)
there appears to be quite a wide range of drain current
that will produce reasonable results. What is most
important is that the current source NOT
supply more current than the JFET will sink - that is,
with the photodiode in complete darkness the
drain-source voltage should be low, giving the
cascode circuit itself several volts of drop in order for
it to work. Practically speaking, this can imply at
least a 4:1 range of current over which the circuit will
function reasonably well.
- Although it bears further analysis to determine optimal
noise performance, what I generally do to set the FET's
drain current (using the bipolar current source) to
determine the current at which the drain voltage starts to
rise suddenly due to excess drain current (again, with the
photodiode in complete darkness) and then "back off" the
drain current by 20%-25% or so. This appears to
provide a "safe" margin of operation over the expected
temperature range.
- Remember: It's best to have the current
source providing too little current (e.g.
low-ish drain-source voltage) than too much!
- The use of metal film resistors throughout.
As compared to standard carbon film resistors, metal film
resistors produce far less shot noise - but in this circuit,
there aren't any resistors located in those portions of
circuits that would likely contribute measurably to the
noise floor so the improvement in performance is likely to
be minimal.
- Cooling of the circuitry. By cooling the
transistors and photodiode, the noise levels will
decrease. From a practical standpoint, one would not
cool the entire circuit, but just the critical parts (e.g.
D1 and Q1 - possibly including C2, R4 and C3.) Note
that cooling components is awkward as it is imperative that
condensation be avoided on the optics and circuitry itself -
not to mention the high current drain and mechanical
vibration if a Peltier
device and fan are used. (Note
that this circuit can be slightly "microphonic", likely owing to minute
changes in parasitic circuit capacitance and component piezoelectrics,
and these mechanical vibrations may be picked up - see additional
comments, below.)
- If your interest is strictly with low-frequency (<200
Hz) operation for modes like WSJT, it is worth reiterating
that zero reverse bias will likely provide optimal
performance because of reduced "1/F" noise which is a phenomenon
that is most dominant at very low frequencies but this
improvement in performance at very low frequencies will come
at the expense of higher frequency (>200 Hz)
performance. Note that the "200 Hz" frequency is
dependent on the capacitance of the photodiodes being used
and that these numbers represent those typically obtained
with a BPW34 in a circuit similar to that shown in Figure
1. This "knee" frequency will be higher with
smaller area photodiodes and lower with larger area
photodiodes. If you wish to do experimentation
at low frequencies, it is recommended that the audio be
tapped at Pin 1 of U1 (in Figure 7) since this is before
any high-frequency emphasis: Note that audio at this
point will not be suitable for voice communications
however as it will sound very muffled. If zero bias
is used, you may need to tweak R5 to reduce the Q1's
current in a usable operating range.
- As with the previous circuit, the value of the
bypass capacitor in the cascode amplifier (in this case,
C4 - 2.2uF) should be chosen with care: If very
large values are used, low-frequency oscillation or
"ringing" may occur - particularly at higher supply
voltages (e.g. >12 volts) or with large,
rapid/transient changes in light.
- Although not part of the circuitry, it is very
important to remember that noiseless gain
can be achieved through the use of larger lenses!
Other Comments:
Improving high (and low) frequency response:
- For more discussion on the use at lower
frequencies (<300 Hz) or with higher-frequency
subcarriers, refer
to this page.
Selection of JFETS
- JFETs must be used instead of insulated-gate FETs (like
MOSFETs) mainly because the gate insulator is a potential
source of noise due to electron migration through that
insulator. Also, the circuit in Figure 5
relies on the fact that the gate-source junction will go
into conduction and establish a reverse bias across the
photodiode in order to work - and this simply would not
happen with an insulated-gate device! While one could
add additional components to a MOSFET circuit to allow such
a circuit to work, keep in mind that those additional
components would also be sources of noise and would likely
degrade the circuit's performance, anyway.
- While low-noise GaAsFET microwave devices may seem to be
reasonable choices for these circuits, many of these these
devices tend to have far higher low-frequency noise than
even inexpensive JFETs - plus they are rather expensive and
fragile by comparison.
- It is possible that this sort of circuit could work with a
bipolar transistor, but thermal stability considerations
(the current gain of a bipolar device is strongly dependent
on temperature) plus the fact that many bipolar transistor
are at their quietest at fairly low collector currents make
a JFET much more attractive than a bipolar device in this
case - but this may be worth experimentation.
- Although this receiver circuit has been used over quite a
wide temperature range (from 0C to over 25C) it is probable
that minor modifications would help to maintain maximum
noise performance over such an operating range. Keep
in mind, of course, that with other things being equal,
noise performance will generally improve with lower
temperature.
As mentioned above, I chose to use a 2N5457 instead of an
MPF102. While the MPF102 is a pretty good device, a quick
glance at the spec sheets will show that it is broadly
characterized - that is, given a hundred devices from different
manufacturers made at different times, you'd see that the
measured parameters were all over the place. The 2N5457 is
a much more consistent device and one is likely to be more
similar to another than MPF102s are to each other. Having
said that, it is still reasonable to obtain many more devices
than you need and sort through them, using only those that have
the best performance. If all you have is a bunch of
MPF102s, it may be worth going through several of them, finding
the one(s) that have the lowest noise - something that is also
likely to be related to the highest zero-bias drain current.
Finally, it is worth mentioning that some JFETs may
NOT
operate in a useful way when a
positive gate voltage is
present. It seems as though many common devices like the
MPF102 and 2N5457 are useful as they continue to provide lower
channel resistance even as the gate goes into conduction as some
"pinching room" of the channel still seems to be available.
There are other JFETs such as the Philips BF862: This JFET
is quite a remarkable device in that its designers seem to have
achieved high transconductance and saturation high current
without inordinately high gate capacitance. To be used
with this circuit, however, modifications will be required as
this FET's drain maximum current is
much higher than that of the 2N5457 - in the
range of 15-25 milliamps. In preliminary testing with this
transistor, the source resistor was removed and the current
source (Q3) was reworked to use a high-beta PNP device (although
another current source topology - such as a
Widlar
or
cascode
current source may be more appropriate) to handle the
current. Initial testing shows that this device is a bit
more "finicky" than the 2N5457 but that its performance it was
at least equal - even if operated at about the same current
(around 3mA) as one would operate a 2N5457: It has yet to
be determined if the much higher drain current capability of
this device will provide any significant advantage of noise.
Good Through-hole FETs
are getting difficult to find these days...
If you are an avid builder, you have probably noticed
that over the past several years, through-hole JFET
transistors have almost disappeared from the catalogs of
manufacturers current offerings! You used
to be able to buy the venerable MPF102 from any number
of places for a low cost, made by a lot of different
manufacturers, but nowadays, you have to resort to
surplus places or even EvilBay. The "problem" with
Ebay is that unless you buy from a reputable vendor, you
are never quite sure what you are getting - even if the
parts that arrive have a particular number stamped on
them!
Such is the case for the transistor described here, the
2N5457 as this, too, has disappeared from the
catalog in the through-hole version - but it is still
available in the surface mount
version: Look for the MMBF5457 (the
suffix will vary).
Don't let the idea of a surface-mount part scare
you: Just use a good pair of glasses, a
fine-tipped soldering iron and some tweezers - and buy a
few extra transistors.
|
Using
Avalanche PhotoDiodes (APDs):
A similar circuit was built using APDs (Avalanche PhotoDiodes)
and is
described
on this page.
Beware of microphonics and current loops!
It is also worth mentioning that, for a number of reasons, that
all of the circuits shown on this page tend to be
somewhat microphonic - that is, they will respond (in differing
degrees) to mechanical vibrations.
It is very
important that any loudspeakers used be located away from the
optical receiver to avoid acoustic feedback! This
simple fact precludes the inclusion of a speaker contained
within the same housing as the receiver itself.
It is again reiterated that it is best that the optical
receivers
NOT share the same power supplies as
either the transmitter or speaker amplifier: Doing so is
inviting trouble, as circulating currents from these other
devices tend to find their way into the (extremely!) sensitive
receiver and will likely result in crosstalk and/or
feedback! It is for this simple reason that the optical
receiver itself has been designed to operate from a single 9
volt battery!
Additional comments about high-sensitivity
optical receivers in general:
Why not use low-noise op amps in the front end?
One might ask why discrete transistors were used instead of
high-performance, low-noise op amps
(like the LT1115,
LMH6624, LMV751 to mention but a few) in the first stage
of the optical detector? The answer is that readily
available op amps - even very good, low-noise ones - will not
perform as well as a single JFET amplifier. Why might this
be? As Bob Pease points out in his article on
Transimpedance Amplifiers (
see the article "What's
All This Transimpedance Amplifier Stuff, Anyway" in the
January 8, 2001 issue of Electronic Design, via the web archive) one has to add
a JFET in front of an op amp in order to obtain the best
possible noise performance for several reasons:
- Input FETs on op amps don't run as "rich" (Bob Pease's
term) as you need them to for lowest noise
performance. Most op amps are designed to operate at
very low currents, so their input FET devices also run at
low currents: As was mentioned before, a JFET is often
quietest when running near its saturation current - and
there's really no way to change an on-chip JFET to fix this.
- The K3PGP circuit (Figure 1) and the circuit in Figure
5 have NO OTHER COMPONENTS connected at
the photodiode-gate junction. At such low signal
levels, the addition of any other components
will contribute noise to the circuit. Op amps just
aren't built this way, so they may have other potential
noise sources. Remember that the thermal noise of any
other devices - including on-chip "components" such as
resistors and current sources - will contribute noise.
- Remember: We are relying on the conductivity of the
gate-source junction of the JFET to help establish a reverse
bias across the photodiode. While this could also be
done with a resistor (as in a feedback loop) remember that
adding such a component would also add another source of
noise!
Of course, one could replace Q2 with an op amp to maintain the
cascode configuration, but that would not likely offer any
performance enhancements: If you do, you must keep in mind
that this stage should be self-biasing (like the Q2 circuit) to
accommodate different voltage/current conditions present at the
drain of Q1.
How about those handy photo amplifier ICs with the
built-in photodiode and op-amp?
Also available are a number of devices that have integrated
photodiodes and op amps contained within a transparent package,
such as the TI (formerly Burr-Brown) OPT101, OPT201, OPT212 and
similar. While these components are useful in minimizing
size and component count, experiments by others indicate that
they offer little - if any - performance advantage over a
less-expensive discrete photodiode coupled to a low-noise op-amp
and have performance that is noticeably
inferior
to that of the VK7MJ circuit across the audio range.
Final comments:
It should be stated once again that the goal was to produce
a highly-sensitive radiometric optical detector that was
optimized for speech range (up to about 3 kHz) frequency
response. Additionally, being self-funded hobbyists, there
was the additional goal that such a detector be built - as much
as possible - using inexpensive, readily available,
off-the-shelf components and construction techniques that were
well with the capabilities of the advanced electronic
hobbyist: With the designs outlined above, we believe that
we have largely achieved that goal!
If, on the other hand, the goal is to achieve optimal
weak-signal detection capabilities at very low (sub-speech)
frequencies or higher bandwidths (above speech, including the
use of high-speed data, video, or multiple carriers) then
careful consideration is warranted when deciding whether or not
the methods outlined elsewhere on this page are entirely
appropriate!
This being said, the author of this page is well-aware of the
fact that other technologies can be brought to bear to provide
further improvements in overall system "sensitivity" - including
(but not limited to) the cooling of the electronics and the use
of more-exotic detectors such as Avalanche Photo Diodes (APDs)
and PhotoMultiplier Tubes (PMTs) - but the use of these types of
components, while worth of experimental pursuit, are largely out
of the practical range of the average self-funded hobbyist!
Acknowledgments:
Credit should be given to the
fine work by K3PGP and VK7MJ for setting the groundwork for
these experiments. Also appreciated are comments by
Yves, F1AVY on the Optical
DX Yahoo group concerning various aspects of the
operation of these circuits.
Related pages:
- The K3PGP
Pages:
- A Low Noise PIN Diode Laser Receiver - Part 1
and Part 2
Note that some of these pages may not render
properly on some browsers.
- "Modulated
Light DX Receiver Circuitry" on
the
Modulated
Light
DX page. These pages contain a wealth
of information on related topics.
- F1AVY's
pages - Yves describes many aspects of
detection (and methods of using lasers to generate
signals.) Please note: The
hosting web site of Yves' pages has changed and most
of the pages at the new site are in French - click
here for a Google translation of the main
page into English.
- Photodiode
Front Ends: The Real Story - This is
an article by P.C.D. Hobbs that describes several
techniques to maximize bandwidth of a photodiode
receiver. For an application using this type
of circuit, see the page LED
AM Video link on this site.
- Photodiode
Amplifiers - Turning Light into Electricity
- From National Semiconductor via the web archive, an online seminar
about various aspects of using photodiodes and how to
amplify their output. This page links not only to
some .PDFs of slides and transcripts of the seminar, but
it also has an online video of the original
presentation. Related to this topic is National
Semiconductor's application note AN-1244, also from the web archive, which contains information about this same topic.
- Hamamatsu
Photonics has, on its website, a number of
papers about the "what and how" of many types of optical
components. For more info, look at:
- The Technical
Notes page (web archive). This page describes the
general theory behind the operation of many types of
optical devices, such as photodiodes, photomultiplier
tubes, and many more devices.
- The Application
Notes page (web archive. This page has a number
of articles describing how optical devices are used in
the real world.
- The Hamamatsu Document Libraray
- This links to a search for documents on various topics, but it does
not have a "browseable" list of titles with descriptions that I could
find...
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to the KA7OEI Optical communications Index page.
If you have questions or comments concerning the
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information at
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Keywords:
Lightbeam
communications,
light
beam,
lightbeam,
laser
beam,
modulated
light,
optical
communications,
through-the-air
optical
communications,
FSO communications, Free-Space Optical
communications, LED communications, laser
communications, LED, laser, light-emitting diode,
lens, fresnel, fresnel lens, photodiode,
photomultiplier, PMT, phototransistor, laser tube,
laser diode, high power LED, luxeon, cree,
phlatlight, lumileds, modulator, detector
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