A highly-sensitive optical receiver optimized for speech bandwidth



Abstract

When designing a receiver that is intended to be used for long-distance, optical through-the-air communications, there are some things to be considered that are very different from what might be required in most other situations where detection (and demodulation) of an optical signal is to be done:
 With the above in mind, the optical receiver was designed to meet those needs while, at the same time, being constructed using fairly easily-obtainable components using conventional techniques.

Please Note:
"Why not use (fill in the blank) detectors instead of photodiodes?"

There are a number of other technologies available that may allow even better "weak signal" performance than standard silicon PIN photodiodes - such as Photomultiplier Tubes and Avalanche Photodiodes.  While these technologies exist, the intent was to come up with a design that was both inexpensive to build and easy to replicate.  Having said this, these more-exotic devices (PMTs and APDs) occasionally appear on the surplus market for reasonable prices and experimentation using such devices is certainly encouraged!

Recommended reading

It is highly recommended that one also read these web pages:

Simplified version of the "Version 3" optical receiver
Figure 1:
Simplified version of the "Version 3" optical receiver - see text.
Click on the image for a larger version.
Simplified version of the Version 3 optical receiver

Figure 1 shows the schematic of an optical receiver from the link above.  First, a few comments about the circuit:
Comment:  The circuit depicted in Figure 1 differs from the prototype "Version 3" mentioned on the aforementioned web page only in that is does not include an extra low-pass filter stage, and or an "active decoupler" in the power supply, the latter not having found to be necessary when the receiver was operated from its own, isolated 9 volt battery power source.

This circuit is intended to be operated from its own, single 9-volt battery - which is one of the reasons why the reverse-polarity protection is present in the form of TH1, a self-resetting thermal fuse, and D4, a reverse-polarity protection diode:  It is extremely easy to momentarily connect a 9-volt battery backwards while fumbling in the dark - something that could instantly destroy U1.  In order to maximize the available voltage, a series protection diode was NOT used!

While the use of an LM833 is shown in the diagram, practically any dual op-amp with "reasonably" low-noise specifications may be used, including the venerable TL082/TL072:  A 1458-type op amp was tried, but it was found to be quite noisy.

Note that operating an LM833 (and many other common dual op-amps) from a single 9-volt battery pushes the low voltage specification of this device - which is listed as 10 volts:  Testing has indicated that the LM833 units that we have seen seem to operate reasonably down to at least 7 volts, but this is not a guaranteed specification!  If you are constructing this circuit, keep in mind that there are many other op-amps that offer good performance but can operate from much lower supply voltages, such as the LM4562 or the LMC6482, to name but a few!

In Figure 1, note also the presence of S1, a "gain" switch across U1B.  This switch is optional and is present to accommodate both situations where signals are very strong (such as with shots over just a few miles) and also those in which signals are very weak, close to the noise floor of the receiver.  Of course, this added gain does nothing to improve the actual sensitivity of the receiver, but it can bring a low-level signal up to the range where one's audio amplifier and monitor/recording system can properly operate!

Note also J1:  This jumper (optionally) shorts out R4, the resistor in the source lead of Q1.  This 10 ohm resistor (and C3) is present solely for the purpose of measuring the source current through Q1 and, after final adjustment, may simply be shorted out if desired.

Finally, note that this circuit is designed such that gate current can actually flow through D1 and into Q1!  This is a rather peculiar way to run a JFET, but this quirk allows the establishment of a reverse bias across D1.
Another version of this optical receiver

Ron, K7RJ, has been helpful in the testing of the "Version 3" receiver design for several years, having been involved in many of the short and long-distance communications experiments.  Because of his interest, he has done something that I have not:  Designed and laid out a circuit board for the receiver.

Using an "online" circuit board manufacturing house, he has also had several boards made for these receivers, which have also been tested.  Furthermore, Ron has put together a web page that has documented his efforts and includes information on how you can get boards made.

Ron's web page is here:  http://home.comcast.net/~ronk7rj/detector/

Note that the schematic on Ron's page differs in some very minor ways with the one shown in Figure 1 (above) and that parts designations (e.g. "R1", "C5", etc.) are different, but all of the other comments about the circuitry made on this page apply to his version - with the appropriate parts translations, of course!

Also note that Ron is NOT set up to supply people with boards directly, but this page provides all the information needed to download the design, download the (free!) board layout program, and get you to the point at which you can take your credit card and get some boards (6 of them) made for you and your friends!  Also included on this page is a parts list and a few pictures pertaining to some of the more-important details of the receiver's construction.

Note that Ron's page includes several pictures and practical advice pertaining to how the photodiode is to be mounted:  These techniques are valid no matter what means of construction you end up choosing!

A few words on the setting of Q1's drain current:

With the use of the cascode circuit consisting of Q1 and Q2, signals from the photodiode, D1, are expressed mainly as variations in the drain current of Q1.  In order for this circuit to work, however, it is necessary that the standing current through Q1 be properly set.

Because of the circuit topology, there is likely to be positive gate-source voltage with a small amount of gate-source current flowing.  As you might expect, this fully "turns on" the JFET.  This operating condition also requires some care in the proper selection of the drain current in order for the circuit to work properly!

Q3 is the primary current source for the JFET, Q1, and as such it has a fairly high intrinsic impedance, with the standing current set through the selection of R5, nominally a 120 ohm resistor.  If the current is too high for the particular transistor used for Q1, the gain and performance is greatly reduced.  In general, it is better to run the JFET at a current that is slightly "too low" than slightly "too high" as the penalties for running slightly "low" current (slightly degraded S/N ratio) are less severe.

For the typical 2N5457 from the various sources onhand, an R5 value of 120 ohms seems to be about right, resulting in consistently "good" performance.  If a less-tightly spec'd JFET is used (such as an MPF102) then one may need to do some experimentation to find the "optimal" value for that specific device.  In experimenting, some MPF102's of various manufacturers and vintages were tried and while some devices simply did not work properly with R5 set at 120 ohms, none needed higher values (which correlated with lower drain current) than the 180-220 ohm area.

One property of JFETs is that the more drain current you can run through them, the "quieter" they can become - up to a point - a pheomenon having to do with "bulk current" effects - and because of this, best performance (e.g. best signal-noise ratio) is obtained at higher drain currents.  Again, if you go overboard and run too much drain current, the device is simply saturated and performance suffers.

Another device with which good success has been obtained is the Philips BF862.  This JFET, available only in a surface-mount package, is quite remarkable in its combination of having fairly high transconductance, moderately low gate capactance, and high drain current capabilities.  In testing, it operated well with drain currents ranging from just a couple of milliamps to about 20 milliamps!  Of course, the same advice applies as with other JFETs in terms of optimize performance with the proper selection of drain current.

Optimizing weak-signal performance:

As mentioned above, "good" weak-signal performance may be obtained by setting a nominal Q1 drain current, but if you wish to eke out the very best possible performance (that is, the best signal-noise ratio) one can "tweak" to find the optimal drain current for the specific device being used.

To do this requires a simple test fixture that will provide a small but consistent optical signal and a means of looking at the receiver's output and discerning the signal-noise ratio.  To do this, I constructed a "photon range" in a room in my house that had no windows simply by attaching a normal red LED to the ceiling and placing the receiver being tested on the floor directly below.  The LED was modulated with a square wave at the test frequency, with a consistent (but small) amount of current - just enough to be percieved to be glowing to the dark-adapted eye.

Connected to the receiver, via shielded audio cable, is a laptop PC running the Spectran program.  With this program, one can take measurements of the strength of the audio signal from the receiver's detection of the modulated LED and compare it with the noise floor.

This scheme has been used many times and with repeatable results, but there are a few important factors to consider:
When optimizing a receiver, one would typically set R5 at a fairly high value (say, 220 ohms) and the parallel fixed resistors of decreasing value (typically starting at 1k-2k) noting the measured signal-noise ratio, the value of the added resistor, and the voltage across R5 for each test.  At some point, further increases in Q1's drain current (e.g. lower values at R5) will start causing a decrease in signal-noise ratio.  Note that while a potentiometer may be used for R5, be warned that these devices can be extremely noisy - especially when such low signal levels are involved due to the probability of wiper noise!  If a potentiometer is used, one must always measure the value and substitute it with a fixed resistor and re-check the signal-noise ratio!

Note that "best" performance does not need to coincide with maximum gain - which is why it is absolutely necessary that one is always measuring signal-to-noise ratio!  If, for example, you were to make a circuit change that caused the signal being received to drop by 3dB - but your noise floor dropped by 6 dB - your receiver is actually able to "hear" 3dB better, despite the drop in gain!


A few observations about the receiver's operational characteristics:

Because no feedback is involved in the operation of the photodiode portion of the circuit in Figure 1, the detector has an intrinsic 6dB/octave amplitude rolloff characteristic above a few hundred Hz - the precise "knee" frequency depending on the the capacitance of the photodiode, the JFET and its cascode circuit, and stray circuit capacitances.  To compensate for this rolloff, U1B, a differentiator circuit is used, flattening out the frequency response over the voice bandwidth.

Below the diode's "knee" frequency, however, the differentiator has the effect of acting as a highpass filter, reducing the level of low frequencies (those below 300 Hz or so) but this has little effect on the reproduction of speech.  One advantage that it offers is that the strong fundamental frequencies associated with urban lighting (120 Hz and 100 Hz for 60 and 50 Hz power mains frequencies, respectively) are significantly reduced, offering some degree of prevention of overload of the audio amplification system used to listen to such communications:  Without this filtering, it is possible that when used near populated areas, the "hum" from urban light sources could overdrive the speaker amplifer before a comfortable listening level could be achieved!

The U1B differentiator circuit also has built-in low-pass filtering, notably R10/C7 and R11/C8 in the feedback path.  These components, along with R6 and C6, work together to provide high-frequency rolloff above 6-8 kHz or so - well above the majority of the speech energy.  Were the differentiator not designed to have a reasonably high frequency cutoff frequency it would, in theory, produce a tremendous amount of high-frequency hiss owing to its 6dB/octave boost - something that would actually distract the listener.

Operation under "high light-level" conditions:

As mentioned before, the "knee" frequency above which amplitude rolls off at about 6dB/octave is typically in the 200-300 Hz range under dark conditions using the BPW34 photodiode.  Because this "knee" frequency is largely a result of the capacitance of the photodiode itself (along with other circuit capacitance, such as the FET, wiring, etc.) in association with various shunt resistances associated with the circuit (e.g. diode and FET leakage, etc.) it should come as no surprise that as light levels increase, this "knee" frequency would change as well!  This effect, caused by increased conductivity of the photodiode itself due to the presence of light, reduces the effective resistance of the circuit, raising the "knee" frequency even higher.

This effect results in the increase of the frequency range over which the frequency response of the "front end" (that portion prior to U1) is flat - that is, it pushes the "knee" frequency upwards.  Since U1B is a differentiator effecting a 6dB/octave boost, as the "knee" frequency is pushed farther into the middle of the speech range, the more "tinny" the audio will sound!

Practically speaking, this effect has been most obvious when running short-distance paths (under 10 miles or 16 kilomters) using a 3-watt LED transmitter:  One could modify the circuit to accommodate such shifts in the "knee" frequency (by modifying the R/C constants associated with U1B), but at such short distances it is usually more convenient to either reduce the power of the distant transmitter, or put an optical attenuator (such as a piece of cardboard with a bunch of holes in it) in front of the receiver's objective lens to reduce the amount of energy reaching the photodiode.

Comment about the use of a lower-capacitance photodiode at D1:

If a lower-capacitance photodiode than the BPW34 is chosen by the builder of an optical receiver, it should be noted that the "knee" frequency will likely be higher than the 200-300 Hz figure mentioned above.  If one wishes to maintain a relatively "flat" frequency response over the speech range, it will be necessary to ascertain the new "knee" frequency of circuit with the chosen photodiode and modify the differentiator circuit (U1B) accordingly!

Noise and frequency response of the circuit:

As one might expect, the limiting factor of the sensitivity of this circuit is the noise intrinsic to the components being used - namely, D1 and Q1.  As much as practical, these sources of noise are minimized, but it is not possible to eliminate them entirely.

For the circuit in Figure 1, the differentiator limits the frequency response at the low end (below about 300 Hz) - but if one were to look at the "pre-differentiator" audio (e.g. the output of U1A) one would see that at very low frequencies (below a few 10's of Hz) that the noise would increase - largely as a result of the inevitable "inverse frequency" noise sources associated with any electronic device:   In the circuit, these noise sources are made somewhat worse by the fact that the photodiode (D1) is reverse-biased, but since the intent was to optimize performance in the speech frequency range (roughly 300-3000 Hz) this is of relatively little importance.

Looking at the "pre-differentiator" audio (the output of U1A) across the speech range, one notes the expected 6dB/octave amplitude rolloff of detected signals.  If one looks at the noise floor, however, one sees that it rolls off, too, but not as fast.  What this means is that given a constant input of an optical signal, the signal-noise ratio will decrease as the frequency increases.  With the addition of the differentiator (U1B) the frequency response is constant, but instead of the level of our test signal decreasing, the noise floor will increase with frequency - eventually reaching a point at which the noise energy is equal to or greater than that of the test signal!

A "perfect" differentiator would have a 6dB/octave increase in gain up to infinity - which, of course is impossible, not to mention undesirable - but since we are interested in speech response, the differentiator's response is intentionally limited to a "reasonable" frequency above the speech range - specifically, around 6-7 kHz with the selected values of R9 and C6.  To further reduce "noise fatigue" of the listener - and to better-remove high-frequency components of, say, a PWM-based transmitter, additional low-pass filtering is provided with the addition of C7/C8.

Use at very low frequencies (<200 Hz):

It is important to note that the circuit in Figure 1 is not intended for very-low (<200 Hz) operation as might be encountered for extremely low-bandwidth, slow signaling techniques such as those that might be encountered using QRSS, WSJT, or similar modes.  Here are a few comments on why this might be:
If this is done, note that higher-frequency (speech!) response will suffer even more.  Also note that one might want to re-check Q1's drain current (doing so with very low test frequencies) to verify optimal signal/noise performance, especially since this can interact with adjustments of R4's value.  Finally, be aware that such a configuration is far more sensitive to ambient light and even very small levels can completely saturate the detector!

If one's interests are mainly frequencies below about 200 Hz, the use of the K3PGP receiver (and its variants) is recommended - see this page for more information and link.


Constructing the circuit:

As simple as the circuit is, there are a few key points that must be made pertaining to its construction:
Additional circuit notes

Operational voltage range:

With the components shown in Figure 1, the circuit is designed to operate properly from about 8 volts to 12 volts.  At and below 7.5 volts, the receiver performance begins to degrade, so one should keep spare 9 volt batteries onhand!

When the supply voltage rises above 12-13 volts, instability arises in the cascode circuit:  If one anticipated operating exclusively at such voltages, one could modify the cascode stage accordingly, or simply add a regulator to keep it in the 9-10 volt range!  Note that only the "front end" portion of the circuit (Q1, Q2, Q3) would need to be regulated in such cases, as the op amp (U1) will happily operate at these higher voltage!

Remember:

Miscellaneous comments on the receiver design:

(Note:  An attempted minimally-technical description follows...)

Because the goal was simply to allow reception of speech, optimal performance was desired in the 300-3000 Hz range.  With this in mind, there are several observations that one might make:
As it turns out, the above two points actually have something to do with each other:  One of the main limiters in the upper frequency response of an optical receiver is the intrinsic frequency response of the detector being used.  Let us take, as an example, the BPW34 PIN photodiode.  This is an inexpensive device ($1.00 US or less in small quantities) that offers reasonably good performance.  It is also a "medium-area" photodiode - that is, it is neither very small or very large, having an active surface area on the order of 7.5 square millimeters.

One unchangeable fact is that the larger the diode, the more self-capacitance that it exhibits.  In the case of the BPW34 at zero volts, it has a capacitance of around 70pF.  While this amount of capacitance may not sound like much - especially at audio frequencies - one must appreciate the very high impedances involved.  For example, if the circuit's impedance were about 10 megohms, one could reasonably expect that frequencies above about 225 Hz would be attenuated at 6dB per octave.

When one is trying to detect extremely weak signals, the primary limitation of the detection system is likely to be noise:  If the level of the signal being sought is below that of the intrinsic noise of the receiver itself, it will be lost!  It makes sense, then, that in a receiver system, that one try to preserve as much of the original signal as possible.  As can be seen from the above, by the time we reach 225 Hz, the signals impinging on the photodiode are already being attenuated and dropping toward the noise floor!

One way to reduce this attenuation is to decrease the capacitance of the photodiode.  Aside from simply using a smaller photodiode, one can also put reverse bias on the diode to exploit a property in which the capacitance decreases.  For example, the same BPW34 diode that has a capacitance of about 70pF at zero volts will have a capacitance of about 25pF at just 3 volts and dropping to about 10pF at 25 volts.  Taking the example above, this would raise the "knee" frequency from 225 Hz at zero volts bias to about 625Hz at 3 volts and about 1600 Hz at 25 volts.  Ideally, going from 0 volts bias to 25 volts would improve the level of the recovered signal at, say, 1000 Hz by roughly 15dB.  (Note that the above example is illustrative only and doesn't take into account other sources of capacitance, such as wiring and the JFET itself.)

The caveat to increasing the bias voltage on the photodiode is that leakage currents also increase - which also results in increased noise!  Fortunately, the majority of this increase in noise occurs at low frequencies, mostly below 300 Hz, so it has only minimal impact on the higher (speech) frequencies.  For the most part, the increase in the diode's noise caused by the higher bias voltage is much less than the amount of signal gained by the reduction of the device's capacitance!

Why not use a feedback amplifier?

Typically, a TransImpedance Amplifier (TIA) is used in conjunction with a photodiode for detection of weak signals.  When properly designed, this offers flat frequency response up to a "knee" frequency - which is dependant, in large part, to the amount of feedback being applied and how it relates to the photodiode's capacitance.  In this way, the effects of the diode's capacitance on frequency response can be greatly mitigated, as it is the properties of the photodiode and the amplifier's closed loop that establish the available bandwidth.

While a TIA is an attractive option, it has the downside of requiring the addition of additional components in the signal path - each component being a potential source of noise.  For example, a TIA circuit has a feedback resistor through which current flows to close the loop - but not only does the current flowing through this resistor produce noise on its own, but noise produced by the amplifier circuit itself (and its associated components) is also fed back into the input, further contributing to signal degradation.

Generally, when building a TIA one uses a low-noise operational amplifier.  As it turns out, however, even specialized "low noise" op amps can be noisier in this situation than a discrete junction FET device!  For this reason, it is not uncommon to see extremely low-noise TIA circuits that use low-noise op amps preceded by more garden-variety FETs - or even ones that avoid the use of op amps entirely.


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Keywords:  Lightbeam communications, light beam, lightbeam, laser beam, modulated light, optical communications, through-the-air optical communications, FSO communications, Free-Space Optical communications, LED communications, laser communications, LED, laser, light-emitting diode, lens, fresnel, fresnel lens, photodiode, photomultiplier, PMT, phototransistor, laser tube, laser diode, high power LED, luxeon, cree, phlatlight, lumileds, modulator, detector
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